Methods of quantizing signals using variable reference signals

ABSTRACT

Methods for reading a data location coupled to an electrical conductor. A counter receives a signal from an analog-to-digital converter coupled to the electrical conductor. The counter produces two or more counts, and in some embodiments, the counts are based in part on a variable reference voltage. An interfuser may be coupled to an output of the counter. The interfuser receives the two or more counts from the counter and reads data conveyed by the data location based on the two or more counts.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a divisional of U.S. patent application Ser. No. 11/818,995, filed on Jun. 15, 2007.

BACKGROUND

1. Field of Invention

Embodiments of the present invention relate generally to devices and, more specifically, in a particular embodiment, to quantizing circuits with variable reference signals.

2. Description of Related Art

Generally, memory devices include an array of memory elements and associated sense amplifiers. The memory elements store data, and the sense amplifiers read the data from the memory elements. To read data, for example, a current is passed through the memory element, and the current or a resulting voltage is measured by the sense amplifier. Conventionally, the sense amplifier measures the current or voltage by comparing it to a reference current or voltage. Depending on whether the current or voltage is greater than the reference, the sense amplifier outputs a value of one or zero. That is, the sense amplifier quantizes the analog signal from the memory element into one of two logic states.

Many types of memory elements are capable of assuming more than just two states. For example, some memory elements are capable of multi-bit (e.g., more than two state) storage. For instance, rather than outputting either a high or low voltage, the memory element may output four or eight different voltage levels, each level corresponding to a different data value. However, conventional sense amplifiers often fail to distinguish accurately between the additional levels because the difference between the levels (e.g., a voltage difference) in a multi-bit memory element is often smaller than the difference between the levels in a single-bit (i.e., two state) memory element. Thus, conventional sense amplifiers often cannot read multi-bit memory elements. This problem may be increased as high performance multi-bit memory elements become increasingly dense, thereby reducing the size of the memory elements and the difference between the levels (e.g., voltage) to be sensed by the sense amplifiers.

A variety of factors may tend to prevent the sense amplifier from discerning small differences in the levels of a multi-bit memory element. For instance, noise in the power supply, ground, and reference voltage may cause an inaccurate reading of the memory element. The noise may have a variety of sources, such as temperature variations, parasitic signals, data dependent effects, and manufacturing process variations. This susceptibility to noise often leads a designer to reduce the number of readable states of the memory element, which tends to reduce memory density and increase the cost of memory.

Conventional sense amplifiers present similar problems in imaging devices. In these devices, an array of light sensors output a current or voltage in response to light impinging upon the sensor. The magnitude of the current or voltage typically depends upon the intensity of the light. Thus, the capacity of the sense amplifier to accurately convert the current or voltage into a digital signal may determine, in part, the fidelity of the captured image. Consequently, noise affecting the sense amplifier may diminish the performance of imaging devices.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 illustrates an electronic device in accordance with an embodiment of the present invention;

FIG. 2 illustrates a memory device in accordance with an embodiment of the present invention;

FIG. 3 illustrates a memory array in accordance with an embodiment of the present invention;

FIG. 4 illustrates a memory element in accordance with an embodiment of the present invention;

FIG. 5 illustrates I-V traces of memory elements storing different values, in accordance with an embodiment of the present invention;

FIG. 6 illustrates noise in the bit-line current during a read operation;

FIG. 7 illustrates a quantizing circuit in accordance with an embodiment of the present invention;

FIG. 8 illustrates a delta-sigma sensing circuit in accordance with an embodiment of the present invention;

FIGS. 9 and 10 illustrate current flow during operation of the quantizing circuit of FIG. 8;

FIGS. 11-13 illustrate voltages in the quantizing circuit of FIG. 8 when sensing small, medium, and large currents, respectively;

FIG. 14 is a graph of bit-line current versus counter output for the quantizing circuit of FIG. 8;

FIG. 15 is a graph of count versus quantizing circuit output in accordance with an embodiment of the present invention;

FIG. 16 illustrates an example of a delta-sigma modulator with a variable reference signal in accordance with an embodiment of the present invention;

FIG. 17 illustrates voltages in the delta-sigma modulator of FIG. 16;

FIG. 18 illustrates I-V traces of memory elements storing different values in accordance with an embodiment of the present invention;

FIG. 19 is a flowchart illustrating the operation of the delta-sigma modulator of FIG. 16; and

FIG. 20 illustrates an example of a system that includes the electronic device of FIG. 2.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

Various embodiments of the present invention are described below. In an effort to provide a concise description of these embodiments, not all features of an actual implementation are described in the specification. It should be appreciated that in the development of any such actual implementation, as in any engineering or design project, numerous implementation-specific decisions must be made to achieve the developers' specific goals, such as compliance with system-related and business-related constraints, which may vary from one implementation to another. Moreover, it should be appreciated that such a development effort might be complex and time consuming but would nevertheless be a routine undertaking of design, fabrication, and manufacture for those of ordinary skill having the benefit of this disclosure.

Some of the subsequently described embodiments may address one or more of the problems with conventional sense amplifiers discussed above. Some embodiments include a quantizing circuit configured to detect small differences in voltages and/or currents. As explained below, the quantizing circuit may sample the measured electrical parameter on multiple occasions and filter, e.g., average or sum, the samples to reduce the impact of noise. As a result, in some embodiments, the quantizing circuit may resolve small differences between voltage or current levels in multi-bit memory elements and/or light sensors, which may allow circuit designers to increase the number of bits stored per memory element and/or the sensitivity of an imaging device. Prior to describing these embodiments and their advantages, the environment in which they may operate is described.

FIG. 1 depicts an electronic device 10 that may be fabricated and configured in accordance with one or more of the present embodiments. The illustrated electronic device 10 includes a memory device 12 that, as explained further below, may include multi-bit memory elements and quantizing circuits. Alternatively, or additionally, the electronic device 10 may include an imaging device 13 having the quantizing circuits.

Myriad devices may embody one or more of the present techniques. For example, the electronic device 10 may be a storage device, a communications device, an entertainment device, an imaging system, or a computer system, such as a personal computer, a server, a mainframe, a tablet computer, a palm-top computer, or a laptop.

FIG. 2 depicts a block diagram of an embodiment of the memory device 12. The illustrated memory device 12 may include a memory array 14, a quantizing circuit 16, a column decoder 18, a column address latch 20, row drivers 22, a row decoder 24, row address latches 26, and control circuitry 28. As described below with reference to FIG. 3, the memory array 14 may include a matrix of memory elements arranged in rows and columns. As will be appreciated, the imaging device 13 (FIG. 1) may include similar features except that in the case of an imaging device 13, the array 14 might comprise an array of imaging elements, such as complementary-metal-oxide semiconductor (CMOS) imaging elements or charge coupled devices (CCDs).

When accessing the memory elements, the control circuitry may receive a command to read from or write to a target memory address. The control circuitry 28 may then convert the target address into a row address and a column address. In the illustrated embodiment, the row address bus 30 transmits the row address to the row address latches 26, and a column address bus 32 transmits column address to the column address latches 20. After an appropriate settling time, a row address strobe (RAS) signal 34 (or other controlling clock signal) may be asserted by the control circuitry 28, and the row address latches 26 may latch the transmitted row address. Similarly, the control circuitry 28 may assert a column address strobe 36, and the column address latches 20 may latch the transmitted column address.

Once row and column addresses are latched, the row decoder 24 may determine which row of the memory array 14 corresponds to the latched row address, and the row drivers 22 may assert a signal on the selected row. Similarly, the column decoder 18 may determine which column of the memory array 14 corresponds with the latched column address, and the quantizing circuit 16 may quantize (e.g., sense) a voltage or current on the selected column. Additional details of reading and writing are described below.

FIG. 3 illustrates an example of a memory array 14. The illustrated memory array 14 includes a plurality of bit-lines 38, 40, 42, 44, and 46 (also referred to as BL0-BL4) and a plurality of word-lines 48, 50, 52, 54, 56, 58, 60, and 62 (also referred to as WL0-WL7). These bit-lines and word-lines are examples of electrical conductors. The memory array 14 further includes a plurality of memory elements 64, each of which may be arranged to intersect one of the bit-lines and one of the word-lines. In other embodiments, imaging elements may be disposed at each of these intersections.

The memory elements and imaging elements may be referred to generally as data locations, i.e., devices or elements configured to convey data, either stored or generated by a sensor, when sensed by a sensing circuit, such as the quantizing circuits discussed below. The data locations may be formed on an integrated semiconductor device (e.g., a device formed on a single crystal of silicon) that also includes the other components of the memory device 12 (or imaging device 13).

In some embodiments, the illustrated memory elements 64 are flash memory devices. The operation of the flash memory elements is described further below with reference to the FIGS. 4 and 5. It should be noted that, in other embodiments, the memory elements 64 may include other types of volatile or nonvolatile memory. For example, the memory elements 64 may include a resistive memory, such as a phase change memory or magnetoresistive memory. In another example, the memory elements 64 may include a capacitor, such as a stacked or trench capacitor. Some types of memory elements 64 may include an access device, such as a transistor or a diode associated with each of the memory elements 64, or the memory elements 64 may not include an access device, for instance in a cross-point array.

FIG. 4 illustrates a circuit 66 that models the operation of an arbitrarily selected memory element 64, which is disposed at the intersection of WL3 and BL0. This circuit 66 includes a capacitor 68, a pre-drain resistor 70 (R_(PD)), a post-source resistor 72 (R_(PS)), and a ground 74. The resistors 70 and 72 model the other devices in series with the memory element 64 being sensed. The illustrated memory element 64 includes a gate 76, a floating gate 78, a drain 80, and a source 82. In the circuit 66, the drain 80 and source 82 are disposed in series between the pre-drain resistor 70 and the post-source resistor 72. The gate 76 is connected to WL3. The pre-drain resistor 70, the drain 80, the source 82, and the post-source resistor 72 are disposed in series on the bit-line BL0. The capacitor 68, which models the capacitance of the bit-line, has one plate connected to ground 74 and another plate connected to the bit-line BL0, in parallel with the memory elements 64.

Several of the components of the circuit 66 represent phenomenon affecting the memory elements 64 when it is sensed. The pre-drain resistor 70 generally represents the drain-to-bitline resistance of the memory elements 64 connected to the bit-line above (i.e., up current from) WL3 when these memory elements 64 are turned on, (e.g., during a read operation). Similarly, the post source resistor 72 generally corresponds to the source-to-ground resistance of the memory elements 64 connected to the bit-line below WL3 when the memory element 64 is sensed. The circuit 66 models electrical phenomena associated with reading the memory elements 64 at the intersection of WL3 and BL0.

The operation of the memory elements 64 will now be briefly described with reference to FIGS. 4 and 5. FIG. 5 illustrates one potential relationship between the bit-line current (I_(BIT)), the word-line voltage (V_(WL)), and the voltage of the floating gate 78 (V_(FG)). As illustrated by FIG. 5, V_(FG) affects the response of the memory element 64 to a given V_(WL). Decreasing the voltage of the floating gate shifts the I-V curve of the memory elements 64 to the right. That is, the relationship between the bit-line current and a word-line voltage depends on the voltage of the floating gate 78. The memory elements 64 may store data by exploiting this effect.

To write data to the memory elements 64, a charge corresponding to the data may be stored on the floating gate 78. The charge of the floating gate 78 may be modified by applying voltages to the source 82, drain 80, and/or gate 76 such that the resulting electric fields produce phenomenon like Fowler-Northam tunneling and/or hot-electron injection near the floating gate 78. Initially, the memory elements 64 may be erased by applying a word-line voltage designed to drive electrons off of the floating gate 78. In some embodiments, an entire column or block of memory elements 64 may be erased generally simultaneously. Once the memory elements 64 are erased, the gate 76 voltage may be manipulated to drive a charge onto the floating gate 78 that is indicative of a data value. After the write operation ends, the stored charge may remain on the floating gate 78 (i.e., the memory elements 64 may store data in a nonvolatile fashion).

As illustrated by FIG. 5, the value stored by the memory element 64 may be read by applying a voltage, V_(WL), to the gate 76 and quantizing (e.g., categorizing) a resulting bit-line current, I_(BIT). Each of the I-V traces depicted by FIG. 5 correspond to a different charge stored on the floating gate, V_(FG), which should not be confused with the voltage that is applied to the gate, V_(WL). The difference in floating gate 70 voltage, V_(FG), between each I-V trace is an arbitrarily selected scaling factor “x.” The illustrated I-V traces correspond to eight-different data values stored by the memory element 64, with a V_(FG) of 0x representing a binary data value of 000, a V_(FG) of 1x representing a binary data value of 001, and so on through V_(FG) of 7x, which represents a binary data value of 111. Thus, by applying a voltage to the gate 76 and measuring the resulting bit-line current, the charge stored on the floating gate 78 may be sensed, and the stored data may be read.

The accuracy with which the bit-line current is quantized may affect the amount of data that a designer attempts to store in each memory element 64. For example, in a system with a low sensitivity, a single bit may be stored on each memory element 64. In such a system, a floating gate voltage V_(FG) of 0x may represent a binary value of 0, and a floating gate voltage V_(FG) of −7x may represent a binary value of one. Thus, the difference in floating gate voltages V_(FG) corresponding to different data values may be relatively large, and the resulting differences and bit-line currents for different data values may also be relatively large. As a result, even low-sensitivity sensing circuitry may quantize (e.g., discern) these large differences in bit-line current during a read operation. In contrast, high-sensitivity sensing circuitry may facilitate storing more data in each memory element 64. For instance, if the sensing circuitry can distinguish between the eight different I-V traces depicted by FIG. 5, then the memory elements 64 may store three bits. That is, each of the eight different charges stored on the floating gate 78 may represent a different three-bit value: 000, 001, 010, 011, 100, 101, 110, or 111. Thus, circuitry that precisely quantizes the bit-line current I_(BIT) may allow a designer to increase the amount of data stored in each memory element 64.

However, as mentioned above, a variety of effects may interfere with accurate measurement of the bit-line current. For instance, the position of the memory elements 64 along a bit-line may affect R_(PD) and R_(PS), which may affect the relationship between the word-line voltage V_(WL) and the bit-line current I_(BIT). To illustrate these effects, FIG. 6 depicts noise on the bit-line while reading from the memory element 64. As illustrated, noise in the bit-line current I_(BIT) may cause the bit-line current I_(BIT) to fluctuate. Occasionally, the fluctuation may be large enough to cause the bit-line current I_(BIT) to reach a level that represents a different stored data value, which could cause the wrong value to be read from the memory elements 64. For instance, if the bit-line current is sensed at time 84, corresponding to an arbitrarily selected peak, a data value of 100 may be read rather than the correct data value of 011. Similarly, if the bit-line current is sensed at time 86, corresponding to an arbitrarily selected local minimum, a data value of 010 may be read rather than a data value of 011. Thus, noise on the bit-line may cause erroneous readings from memory elements 64.

FIG. 7 depicts a quantizing circuit 16 that may tend to reduce the likelihood of an erroneous reading. The illustrated quantizing circuit 16 includes an analog-to-digital converter 88 and a digital filter 90 connected to each of the bit-lines 38, 40, 42, 44, and 46, respectively. Each bit-line 38, 40, 42, 44, and 46 may connect to a different analog-to-digital converter 88 and digital filter 90. The digital filters 90, in turn, may connect to an input/output bus 92, which may connect to a column decoder 18, a column address latch 20, and/or control circuitry 28 (see FIG. 2).

In operation, the quantizing circuit 16 may quantize (e.g., digitize) analog signals from the memory elements 64 in a manner that is relatively robust to noise. As explained below, the quantizing circuit 16 may do this by converting the analog signals into a bit-stream and digitally filtering high-frequency components from the bit-stream.

The analog-to-digital converter 88 may be a one-bit, analog-to-digital converter or a multi-bit, analog-to-digital converter. In the present embodiment, an analog-to-digital converter 88 receives an analog signal from the memory element 64, e.g., a bit-line current I_(BIT) or a bit-line voltage V_(BL), and outputs a bit-stream that represents the analog signal. The bit-stream may be a one-bit, serial signal with a time-averaged value that generally represents the time-averaged value of the analog signal from the memory element 64. That is, the bit-stream may fluctuate between values of zero and one, but its average value, over a sufficiently large period of time, may be proportional to the average value of the analog signal from the memory element 64. In certain embodiments, the bit-stream from the analog-to-digital converter 88 may be a pulse-density modulated (PDM) version of the analog signal. The analog-to-digital converter 88 may transmit the bit-stream to the digital filter 90 on a bit-stream signal path 94.

The digital filter 90 may digitally filter high-frequency noise from the bit-stream. To this end, the digital filter 90 may be a low-pass filter, such as a counter, configured to average (e.g., integrate and divide by the sensing time) the bit-stream over a sensing time, i.e., the time period over which the memory element 64 is read. (Alternatively, in some embodiments, the digital filter 90 is configured to integrate the bit-stream without dividing by the sensing time.) As a result, the digital filter 90 may output a value that is representative of both the average value of the bit-stream and the average value of the analog signal from the memory element 64. In some embodiments, the digital filter 90 is a counter, and the cut-off frequency of the digital filter 90 may be selected by adjusting the duration of the sensing time. In the present embodiment, increasing the sensing time will lower the cutoff frequency. That is, the frequency response of the digital filter 90 may be modified by adjusting the period of time over which the bit-stream is integrated and/or averaged before outputting a final value. The frequency response of the digital filter 90 is described further below with reference to FIG. 15. For multi-bit memory elements 64, the output from the digital filter 90 may be a multi-bit binary signal, e.g., a digital word that is transmitted serially and/or in parallel.

Advantageously, in certain embodiments, the quantizing circuit 16 may facilitate the use of multi-bit memory elements 64. As described above, in traditional designs, the number of discrete data values that a memory element 64 stores may be limited by sense amps that react to noise. In contrast, the quantizing circuit 16 may be less susceptible to noise, and, as a result, the memory elements 64 may be configured to store additional data. Without the high frequency noise, the intervals between signals representative of different data values may be made smaller, and the number of data values stored by a given memory element 64 may be increased. Thus, beneficially, the quantizing circuit 16 may read memory elements 64 that store several bits of data, e.g., 2, 3, 4, 5, 6, 7, 8, or more bits per memory element 64.

Although the quantizing circuit 16 may sense the signal from the memory element 64 over a longer period of time than conventional designs, the overall speed of the memory device 12 may be improved. As compared to a conventional device, each read or write operation of the memory device 12 may transfer more bits of data into or out of the memory element 64. As a result, while each read or write operation may take longer, more data may be read or written during the operation, thereby improving overall performance. Further, in some memory devices 12, certain processes may be performed in parallel with a read or write operation, thereby further reducing the overall impact of the longer sensing time. For example, in some embodiments, the memory array 14 may be divided into banks that operate at least partially independently, so that, while data is being written or read from one bank, another bank can read or write data in parallel.

FIG. 8 illustrates details of one implementation of the quantizing circuit 16. In this embodiment, the digital filter 90 is a counter, and the analog-to-digital converter 88 is a first-order delta-sigma modulator. The illustrated delta-sigma modulator 88 may include a latched comparator 96, a capacitor 98, and a switch 100. In other embodiments, other types of digital filters and analog-to-digital converters may be employed, such as those described below in reference to FIGS. 17 and 18.

As illustrated, an input of the counter 90 may connect to the bit-stream signal path 94, which may connect to an output of the comparator 96. The output of the comparator 96 may also connect to a gate of the switch 100 by a feedback signal path 102. The output terminal (e.g., source or drain) of the switch 100 may connect in series to one of the bit-lines 38, 40, 42, 44, or 46, and the input terminal of the switch 100 may connect to a reference current source 104 (I_(REF)). One plate of the capacitor 98 may connect to one of the bit-lines 38, 40, 42, 44, or 46, and the other plate of the capacitor 98 may connect to ground.

The illustrated counter 90 counts the number of clock cycles that the bit-stream 94 is at a logic high value or logic low value during the sensing time. The counter may count up or count down, depending on the embodiment. In some embodiments, the counter 90 may do both, counting up one for each clock cycle that the bit-stream has a logic high value and down one for each clock cycle that the bit-stream has a logic low value. Output terminals (D0-D5) of the counter 90 may connect to the input/output bus 92 for transmitting the count. The counter 90 may be configured to be reset to zero or some other value when a reset signal is asserted. In some embodiments, the counter 90 may be a series connection of D-flip flops, e.g., D-flip flops having SRAM or other memory for storing an initial value and/or values to be written to the memory element 64.

In the illustrated embodiment, the clocked comparator 96 compares a reference voltage (V_(REF)) to the voltage of one of the bit-lines 38, 40, 42, 44, or 46 (V_(BL)), which may be generally equal to the voltage of one plate of the capacitor 98. The comparator 96 may be clocked (e.g., falling and/or rising edge triggered), and the comparison may be performed at regular intervals based on the clock signal, e.g., once per clock cycle. Additionally, the comparator 96 may latch, i.e., continue to output, values (V_(FB)) between comparisons. Thus, when the clock signals the comparator 96 to perform a comparison, if V_(BL) is less than V_(REF), then the comparator 96 may latch its output to a logic low value, as described below in reference to FIG. 9. Conversely, if V_(BL) is greater than V_(REF), then the comparator 96 may latch a logic high value on its output, as described below in reference to FIG. 10. As a result, the illustrated comparator 96 outputs a bit-stream that indicates whether V_(BL) is larger than V_(REF), where the indication is updated once per clock cycle.

Advantageously, in some embodiments, the quantizing circuit 16 may include a single comparator (e.g., not more than one) for each column of multi-level memory elements 64. In contrast, conventional sense amplifiers often include multiple comparators to read from a multi-bit memory cell, thereby potentially increasing device complexity and cost.

The capacitor 98 may be formed by capacitive coupling of the bit-lines 38, 40, 42, 44, and 46. In other designs, this type of capacitance is referred to as parasitic capacitance because it often hinders the operation of the device. However, in this embodiment, the capacitor 98 may be used to integrate differences between currents on the bit-lines 38, 40, 42, 44, or 46 and the reference current to form the bit-stream, as explained further below. In some embodiments, the capacitor 98 may be supplemented or replaced with an integrated capacitor that provides greater capacitance than the “parasitic” bit-line capacitance.

The illustrated switch 100 selectively transmits current I_(REF) from the reference current source 104. In various embodiments, the switch 100 may be a PMOS transistor (as illustrated in FIGS. 8-10) or an NMOS transistor (as illustrated in FIG. 17) controlled by the V_(FB) signal on the feedback signal path 102.

The operation of the quantizing circuit 16 will now be described with reference to FIGS. 9-12. Specifically, FIGS. 9 and 10 depict current flows in the quantizing circuit 16 when the comparator 96 is latched low and high, respectively. FIG. 11 illustrates V_(BL), the bit-stream output from the comparator 96, and the corresponding increasing count of the counter 90 for a relatively small bit-line current. FIG. 12 depicts the same voltages when measuring a medium sized bit-line current, and FIG. 13 depicts these voltages when measuring a relatively large bit-line current.

To sense the current through the memory element 64, the illustrated delta-sigma modulator 88 exploits transient effects to output a bit-stream representative of the bit-line current I_(BIT). Specifically, the delta-sigma modulator 88 may repeatedly charge and discharge the capacitor 98 with a current divider that subtracts the bit-line current I_(BIT) from the reference current I_(REF). Consequently, a large current through the memory element 64 may rapidly discharge the capacitor 98, and a small current through the memory element 64 may slowly discharge the capacitor 98.

To charge and discharge the capacitor 98, the delta-sigma modulator 88 switches between two states: the state depicted by FIG. 9 (hereinafter “the charging state”) and the state depicted by FIG. 10 (hereinafter “the discharging state”). Each time the delta-sigma modulator 88 transitions between these states, the bit-stream changes from a logic high value to a logic low value or vice versa. The proportion of time that the delta-sigma modulator 88 is in the state illustrated by either FIG. 9 or FIG. 10 may be proportional to the size of the bit-line current I_(BIT) through the memory element 64. The larger the bit-line current I_(BIT), the more time that the delta-sigma modulator 88 is in the state illustrated by FIG. 9, rather than the state illustrated by FIG. 10, and the more time that the bit-stream has a logic low value.

Starting with the charging state (FIG. 9), the capacitor 98 may initially accumulate a charge (e.g., become more charged). To this end, the output of the comparator 96 is latched to logic low, which, as mentioned above, may occur when V_(BL) is less than V_(REF). The logic low may be conveyed to switch 100 by the feedback signal path 102, and the switch 100 may close, thereby conducting the reference current I_(REF) through one of the bit-lines 38, 40, 42, 44, or 46, as indicated by the larger arrows in FIG. 9. A portion of the electrons flowing through the reference current source 104 may be accumulated by the capacitor 98, as indicated by the smaller-horizontal arrows, and the remainder may be conducted through the memory element 64, i.e., the bit-line current I_(BIT), as indicated by the smaller vertical arrows. Thus, the capacitor 98 may accumulate a charge, and V_(BL) may increase.

The comparator 96 and the reference current source 104 may cooperate to charge the capacitor 98 for a discrete number of clock cycles. That is, when the delta-sigma modulator 88 transitions to the charging state, the delta-sigma modulator 88 may remain in this state for an integer number of clock cycles. In the illustrated embodiment, the comparator 96, the output of which is latched, changes state no more than once per clock cycle, so the switch 100, which is controlled by the output of the comparator 96, V_(FB), conducts current for a discrete number of clock cycles. As a result, the reference current source 104 conducts current I_(REF) through the bit-line and into the capacitor 98 for an integer number of clock cycles.

After each clock cycle of charging the capacitor 98, the delta-sigma modulator 88 may transition from the charging state to the discharging state, which is illustrated by FIG. 10, depending on the relative values of V_(BL) and V_(REF). Once per clock cycle (or at some other appropriate interval, such as twice per clock cycle), the comparator 96 may compare the voltage of the capacitor V_(BL) to the reference voltage V_(REF). If the capacitor 98 has been charged to the point that V_(BL) is greater than V_(REF), then the output of the comparator 96 may transition to logic high, as illustrated in FIG. 10. The logic high signal may be conveyed to the switch 100 by the feedback signal path 102, thereby opening the switch 100. As a result, the reference current source 104 may cease conducting current through the memory element 64 and into the capacitor 98, and the capacitor 98 may begin to discharge through the memory element 64.

In the present embodiment, the delta-sigma modulator 88 discharges the capacitor 98 for a discrete number of clock intervals. After each clock cycle of discharging the capacitor 98, the delta-sigma modulator 88 compares V_(BL) to V_(REF). If V_(BL) is still greater than V_(REF), then the comparator 96 may continue to output a logic high signal, i.e., V_(FB)=1, and the switch 100 remains open. On the other hand, if enough current has flowed out of the capacitor 98 that V_(BL) is less than V_(REF), then the comparator 96 may output a logic low signal, i.e., V_(FB)=0, and the switch 100 may close, thereby transitioning the delta-sigma modulator 88 back to the charging state and initiating a new cycle.

The counter 90 may count the number of clock cycles that the delta-sigma modulator 88 is in either the charging state or the discharging state by monitoring the bit-stream signal path 94. The bit-stream signal path 94 may transition back and forth between logic high and logic low with the output of the comparator 96, V_(FB), and the counter 90 may increment and/or decrement a count once per clock cycle (or other appropriate interval) based on whether the bit-stream is logic high or logic low. After the sensing time has passed, the counter 90 may output a signal indicative of the count on output terminals D0-D5. As explained below, the count may correspond, e.g., proportionally, to the bit-line current, I_(BIT).

FIGS. 11-13 illustrate voltages V_(FB) and V_(BL) in the quantizing circuit 16 when reading data from a memory element 64. Specifically, FIG. 11 illustrates a low-current case, in which the value stored by the memory element 64 is represented by a relatively low bit-line current. Similarly, FIG. 12 illustrates a medium-current case, and FIG. 13 illustrates a high-current case. In each of these figures, the ordinate of the lower trace represents the voltage of the bit-stream signal path 94,V_(FB), and the ordinate of the upper trace illustrates the bit-line voltage, V_(BL). The abscissa in each of the traces represents time, with the lower trace synchronized with the upper trace, and the duration of the time axes is one sensing time 106.

As illustrated by FIG. 11, the counter 90 is initially preset to zero (or some other appropriate value) by applying a reset signal. In some embodiments, the delta-sigma modulator 88 may undergo a number of start-up cycles to reach steady-state operation before initiating the sensing time and resetting the counter 90. At the beginning of the illustrated read operation, the delta-sigma modulator 88 is in the charging state, which charges the capacitor 98 and increases V_(BL), as indicated by dimension arrow 108. At the beginning of the next clock cycle, the comparator 96 compares the bit-line voltage to the reference voltage and determines that the bit-line voltage is greater than the reference voltage. As a result, the bit-stream signal path 94 (V_(FB)) transitions to a logic high voltage, and the delta-sigma modulator 88 transitions to the discharging state. Additionally, the counter 90 increments the count by one to account for one clock cycle of the bit-stream signal 94 holding a logic low value. Next, the charge stored on the capacitor 98 drains out through the memory element 64, and the bit-line voltage drops until the comparator 96 determines that V_(BL) is less than V_(REF), at which point the cycle repeats. The cycle has a period 112, which may be divided into a charging portion 114 and a discharging portion 116. Once during each cycle in the sensing time 106, the count stored in the counter 90 may increase by one. At the end of the sensing time 106, the counter 90 may output the total count.

A comparison of FIG. 11 to FIGS. 12 and 13 illustrates why the count correlates with the bit-line current. In FIG. 13, the high-current case, the stored charge drains from the capacitor 98 quickly, relative to the other cases, because the bit-line current I_(BIT) is large and, as a result, the delta-sigma modulator 88 spends more time in the charging state than the discharging state. As a result, the bit-stream has a logic low value for a large portion of the sensing time 106, thereby increasing the count.

The capacitance of the capacitor 98 may be selected with both the clock frequency and the range of expected bit-line currents in mind. For example, the capacitor 98 may be large enough that the capacitor 98 does not fully discharge (e.g., saturate) when the bit-line current I_(BIT) is either at its lowest expected value or at its highest expected value. That is, in some embodiments, the capacitor 98 generally remains in a transient state while reading the memory element 64. Similarly, the frequency at which the comparator 96 is clocked may affect the design of the capacitor 98. A relatively high frequency clock signal may leave the capacitor 98 with relatively little time to discharge or saturate between clock cycles, thereby leading a designer to choose a smaller capacitor 98.

Similarly, the size of the reference current may be selected with the range of expected bit-line currents in mind. Specifically, in certain embodiments, the reference current is less than the largest expected bit-line current I_(BIT), so that, in the case of maximum bit-line current I_(BIT), the capacitor 98 can draw charge from the reference current while the rest of the reference current flows through the memory element 64.

FIG. 14 illustrates the relationship between the bit-line current I_(BIT) and the count for the presently discussed embodiment. As illustrated by FIG. 14, the count corresponds with (e.g., is generally proportional to) the bit-line current I_(BIT). This relationship is described by the following equation (Equation 1), in which N_(ST) represents the number of clock cycles during the sensing time:

I _(BIT) /I _(REF)=Count/N _(ST)

Thus, in the illustrated embodiment, the count corresponds with (e.g., is indicative of) the bit-line current I_(BIT), which corresponds with the value stored by the memory element 64.

Advantageously, the quantizing circuit 16 may quantize (e.g., categorize) the bit-line current I_(BIT) as falling into one of a large number of categories, each of which is represented by an increment of the count. In doing so, in some embodiments, the quantizing circuit 16 may resolve small differences in the bit-line current I_(BIT). The resolution of the quantizing circuit 16 may be characterized by the following equation (Equation 2), in which I_(MR) represents the smallest resolvable difference in bit-line current I_(BIT), i.e., the resolution of the quantizing circuit 16:

I _(MR) =I _(REF) /N _(ST)

Thus, the resolution of the quantizing circuit 16 may be increased by increasing the sensing time or the clock frequency or by decreasing I_(REF), which may limit the maximum cell current since I_(NIR) is less than I_(REF).

The resolution of the quantizing circuit 16 may facilitate storing multiple bits in the memory element 64 or sensing multiple levels of light intensity in an image sensor element. For example, if the quantizing circuit 16 is configured to quantize (e.g., categorize) the bit-line current I_(BIT) into one of four different levels, then the memory element 64 may store two-bits of data or, if the quantizing circuit 16 is configured to categorize the bit-line current I_(BIT) into one of eight different current levels, then the memory element 64 may store three-bits of data. For the present embodiment, the number of bits stored by the memory element 64 may be characterized by the following equation (Equation 3), in which N_(B) represents the number of bits stored by a memory element 64 and I_(RANGE) represents the range of programmable bit-line currents through the memory element 64:

N _(B)=log(I _(RANGE) /I _(MR))/log 2

In short, in the present embodiment, greater resolution translates into higher density data storage for a given memory element 64.

FIG. 15 is a graph that illustrates one way in which the counter 90 may be configured to further reduce the effects of noise. In FIG. 15, the abscissa represents the count, and the ordinate represents the output of the quantizing circuit 16. In the present embodiment, the three-least-significant digits of the count are disregarded as potentially corrupted by noise. That is, D0-D2 (FIG. 8) either do not connect to the input/output bus 92 or are not interpreted as conveying data that is stored by the memory element 64. As a result, a range of counter values may represent a single data value stored by the memory element 64. For example, in the present embodiment, count values ranging from 00 1000 to 00 1111 are construed as representing a data value of 001. Representing data in this manner may further reduce the effects of noise because, even if noise affects the count, in many embodiments, it would have to affect the count in a consistent manner over a substantial portion of the sensing time to affect the more significant digits of the count. That is, disregarding less significant digits may lower the cutoff frequency of the counter 90. In other embodiments, fewer, more, or no digits may be truncated from the count as potentially representing noise.

Truncating less significant digits may introduce a rounding error, or a downward bias, in the output. This effect may be mitigated by presetting (e.g., driving latches to a particular state in advance of counting or storing a value in memory) the counter 90 in a manner that accounts for this bias. The counter 90 may be preset either before reading from the memory element 64 or before writing to the memory element 64. In some embodiments, the preset value may be one-half of the size of the range of counter values that represent a single output value. In other words, if m digits are truncated from the output, then the counter 90 may be preset to one-half of 2^(m) before reading from a memory element 64 or before writing to the memory element 64. In some embodiments, the memory in the counter 90 may store this preset value.

In some of the previously described embodiments, the reference voltage V_(REF) is generally constant while sensing the data location 64. This is not necessarily the case in the embodiment of FIG. 16, which illustrates an example of a quantizing circuit 120 with a varying reference voltage V_(REF). Quantizing circuit 120 may read data by sensing the data location 64 under changing conditions, e.g., different bit-line voltages V_(BL), which may be altered by changing the reference voltage V_(REF). As described below, the response of the data location 64 to each condition, as well as the change in its response to each change in conditions, may convey information about stored data. Further, this information may be aggregated to improve the precision with which the quantizing circuit 120 writes to, and reads from, the data location 64. Such advantages are described further below, after the components and operation of the quantizing circuit 120 are described.

The illustrated quantizing circuit 120 includes the features of the previously described quantizing circuit 16 (FIG. 8) along with some additional components. Among the additional components are a controller 122 and an interfuser 124, e.g., a component configured to identify a quantity (e.g., a charge on a floating gate) based on multiple measurements indicative of that quantity. Specific examples of how the interfuser 124 may combine measurements are described below.

These additional components 122 and 124 may communicate with other portions of the quantizing circuit 120. In the present embodiment, the controller 122 communicates with three other components: an inverting input of the comparator 96 via a reference signal path 126, an input of the counter 90 via a reset signal path 128, and an input of the interfuser 124 via a state signal path 130. The illustrated interfuser 124 may couple to both the counter 90 and the input/output bus 92 via a plurality of digit signal paths labeled D0-D5. Other embodiments may include more or fewer digit signal paths. In some embodiments, the number of digit signal paths coupling the interfuser to the counter may be different (i.e., less or more) than the number of digit signal paths coupling the interfuser 124 to the input/output bus 92.

Like some of the previously described embodiments, the components of the illustrated quantizing circuit 120 may be formed on an integrated semiconductor device. However, in some embodiments, one or more of the components of the illustrated quantizing circuit 120 may be disposed on another chip or device.

The illustrated controller 122 may be configured to coordinate the operation of the quantizing circuit 120. For example, the controller 122 may vary the reference voltage V_(REF) applied to the inverting input of the comparator 96. In some embodiments, the controller 122 may vary V_(REF) according to a process described below in reference to FIG. 19. When V_(REF) changes, the controller 122 may also signal the interfuser 124 on the state signal path 130, thereby indicating that the quantizing circuit 120 is measuring the data location 64 under a different set of conditions. In certain embodiments, when sending the state change signal, the controller 122 may also transmit a reset signal to the counter 90, which may cause the counter 90 to shift its count to the interfuser 124 and reset the count. Specific sequences of these signals are described below.

The operation of the quantizing circuit 120 will now be described with reference to the following figures: FIG. 17, which illustrates voltages in the quantizing circuit over time; FIG. 18, which illustrates the response of a floating gate transistor to different source-to-drain voltages and floating gate charges; and FIG. 19, which illustrates a process for using data from FIG. 17 to identify a floating gate charge in FIG. 18.

As noted, FIG. 17 illustrates voltages in the quantizing circuit 120 when reading data from the data location 64. Specifically, the top portion of FIG. 17 illustrates the bit-line voltage V_(BL) over time, and the lower trace illustrates the complement of the feedback voltage V_(FB) (i.e., the bit-stream) over time. In the figure, the illustrated time period is divided into two portions: a first sensing time 132 and a second sensing time 134. Between these two time periods 132 and 134, the reference voltage V_(REF) changes from a low reference voltage to a high reference voltage. Within a time period 132 or 134, in this embodiment, the reference voltage is generally constant. The change in the reference voltage V_(REF) may be larger than it appears in the figure. To depict variation in the bit-line voltage V_(BL), the difference between V_(REF-LOW) and V_(REF-HIGH) is compressed.

With the exception of the change in reference voltage V_(REF), the traces illustrated by FIG. 17 are similar to those discussed above with reference to FIGS. 11-13. As previously described, the delta-sigma modulator 88 (FIG. 16) may attempt to keep the bit-line voltage V_(BL) above the reference voltage V_(REF) by elevating the feedback voltage V_(FB) whenever the bit-line voltage V_(BL) is less than the reference voltage V_(REF). As described, elevating the feedback voltage V_(BL) (or its compliment, in some embodiments) turns on the current switch 100, and as a result, the reference current I_(REF) flows into the bit-line, thereby elevating its voltage V_(BL). Thus, the ratio of the sensing time that V_(BL) is high is proportional to the current leaving the bit-line I_(BIT) through the data location 64.

In this embodiment, when V_(REF) increases, the set-point of the delta-sigma modulator 88 changes and V_(BL) rises, thereby driving more current I_(BIT) through the data location 64 to ground 74. As a result, in this embodiment, the count accumulates faster during the second sensing time 134 than during the first sensing time 132, because a higher V_(REF) leads to a higher V_(BL), which drives a larger bit-line current I_(BIT) to ground 74. A larger I_(BIT) may cause V_(FB) to remain at logic high for more time to accommodate the larger bit-line current I_(BIT), and the count may accumulate faster.

The change that results from changing V_(REF) may depend on the type of data location and its state. An example of the relationship between bit-line current I_(BIT) and bit-line voltage V_(BL) is illustrated by FIG. 18. In this embodiment, the data location 64 is a floating gate transistor, so higher source-to-drain voltages V_(BL) may tend to produce larger currents through the data location 64 I_(BIT). This effect may depend, in part, on the charge on the floating gate V_(FG), as indicated by the eight-different traces, corresponding to eight-different floating gate charges, labeled 0x through −7x, where x is an arbitrarily selected scaling constant.

For each of the traces, both the current at a given voltage and the overall shape of the trace may be distinct from the other traces. For example, in the trace corresponding to a floating gate charge of −6x, V_(REF-LOW) produces a bit-line current I₁, and V_(REF-HIGH) produces a bit line current I₂. Not only are both of these currents I₁ and I₂, considered alone, different from the I_(BIT) through transistors with different V_(FG) at the same V_(BL), the relationship between I₁ and I₂ is also different for each V_(FG). Specifically, in this embodiment, the slope of each trace changes depending on V_(FG). Thus, in this embodiment, given the two coordinates represented by I₁ and I₂, each trace has three distinguishing characteristics: the value of I₁, the value of I₂, and the slope from V_(REF-LOW) to V_(REF-HIGH). This additional information about the response of the data location 64 to different voltages can be used to identify the floating gate charge and, thereby, read data.

In some embodiments, the quantizing circuit 120 may read data by performing a sensing process 140 illustrated by FIG. 19. The illustrated process 140 begins with generating a first value by sensing a data location at a first voltage, as illustrated by block 142. In certain embodiments, generating the first value may include holding the data location 64 (FIG. 16) at a first voltage V_(REF-LOW) with the delta-sigma modulator 88 and generating a first count (hereinafter referred to as C₁) based on a bit-stream output from the delta-sigma modulator 88.

After C₁ is generated, the controller 124 may signal the interfuser 124 to latch the output of the counter 90 and, thereby, receive the first value. Thus, the interfuser 124 may include memory. Also, in some embodiments, the controller 122 may reset the counter 90 to prepare the counter to generate a second value under different conditions.

Next in the process 140, a second value is generated by sensing the data location at a second voltage, as illustrated by block 144. The second value may be a count (referred to as C₂) from the delta-sigma modulator 88 (FIG. 16). In some embodiments, the second voltage may be different from the first voltage, e.g., higher, or lower. For instance, the second voltage may be V_(REF-HIGH).

The second value may be generated generally consecutively with, e.g., immediately after, generating the first value. However, in some embodiments, there may be a waiting period between generating the first value and the second value. For example, the controller 122 in the quantizing circuit 120 (FIG. 16) may delay the start of the second count while the delta-sigma modulator 88 reaches steady state operation under the new reference voltage, V_(REF-HIGH). In other embodiments, the second value may be added to the first value, and the counter 90 may continue counting as the reference voltage V_(REF) changes.

After generating the first value and the second value, they may be combined to identify data stored by the data location, as illustrated by block 146 (FIG. 19). Combining these values may include applying the values to a data fusion algorithm, i.e., an algorithm that generates a single output value based on two or more input values. An example is described below in reference to the quantizing circuit 120 of FIG. 16.

After the second count C₂ is generated, the controller 122 may signal the interfuser 124 to latch the outputs of the counter 90 and receive the second count C₂, and this second value may be combined with the first value to identify the data stored by the data location 64. In some embodiments, the interfuser 124 may combine the first value with its second value according to an equation. That is, the data may be a function of C₁ and C₂, which are sensed under different conditions. Below, is an example of such an equation, in which E₁, E₂, and E₃ are empirically or analytically determined constants:

Data=E ₁ ·C ₁ ·+E ₂ ·C ₂ +E ₃(C ₂ −C ₁)/(V _(REF-HIGH) −V _(REFLOW))

The constants E₁, E₂, and E₃ may be determined by testing or modeling the operation of a data location. The constants E₁, E₂, and E₃ may be selected to minimize the likelihood of an erroneous reading. By changing these values, different weights may be attached to different terms depending on its descriptive strength. In some embodiments, V_(REF-HIGH) and V_(REF-LOW) may be generally fixed or constant, and the reciprocal of their difference may be incorporated into the constant E₃.

In other embodiments, a variety of other equations or sensor fusion techniques may be employed to identify the stored data. Examples of other types of sensor fusion algorithms include a Kalman filter, a Bayesian network, or a neural network.

In some embodiments, additional values may be generated by sensing the data location 64 at other voltages. For instance, a third value may be generated at a third voltage, and a fourth value may be generated at a fourth voltage. These additional values may be combined with the first and the second values to identify data stored by the data location.

Other embodiments may read data by applying multiple stimuli to different kinds of data locations. For instance, the data location 64 may be a photo-diode, a CCD device, a CMOS image sensor, a phase change memory, a magneto-resistive memory, or other type of resistive memory.

Advantageously, the illustrated process 140 may combine the separate values to identify the stored data with greater precision than with an individual value. For example, sensing the floating gate transistor characterized in FIG. 18 at two different voltages may provide more information to select among the traces illustrated by FIG. 18 and determine the best-fit profile. Thus, by measuring a profile of the data location 64 (i.e., the response of the data location to a stimulus under changing conditions), the state of the data location can be identified with greater fidelity, and the stored data can be read more precisely.

FIG. 20 depicts an exemplary processor-based system 310 that includes the memory device 12 (FIG. 2). Alternatively or additionally, the system 310 may include the imaging device 13. The system 310 may be any of a variety of types such as a computer, pager, cellular phone, personal organizer, control circuit, etc. In a typical processor-based system, one or more processors 312, such as a microprocessor, control the processing of system functions and requests in the system 310. The processor 312 and other subcomponents of the system 310 may include quantizing circuits, such as those discussed above.

The system 310 typically includes a power supply 314. For instance, if the system 310 is a portable system, the power supply 314 may advantageously include a fuel cell, permanent batteries, replaceable batteries, and/or rechargeable batteries. The power supply 314 may also include an AC adapter, so the system 310 may be plugged into a wall outlet, for instance. The power supply 314 may also include a DC adapter such that the system 310 may be plugged into a vehicle cigarette lighter, for instance.

Various other devices may be coupled to the processor 312 depending on the functions that the system 310 performs. For instance, a user interface 316 may be coupled to the processor 312. The user interface 316 may include buttons, switches, a keyboard, a light pen, a mouse, a digitizer and stylus, and/or a voice recognition system, for instance. A display 318 may also be coupled to the processor 312. The display 318 may include an LCD, an SED display, a CRT display, a DLP display, a plasma display, an OLED display, LEDs, and/or an audio display, for example. Furthermore, an RF sub-system/baseband processor 320 may also be coupled to the processor 312. The RF sub-system/baseband processor 320 may include an antenna that is coupled to an RF receiver and to an RF transmitter (not shown). One or more communication ports 322 may also be coupled to the processor 312. The communication port 322 may be adapted to be coupled to one or more peripheral devices 324 such as a modem, a printer, a computer, or to a network, such as a local area network, remote area network, intranet, or the Internet, for instance.

The processor 312 generally controls the system 310 by implementing software programs stored in the memory. The memory is operably coupled to the processor 312 to store and facilitate execution of various programs. For instance, the processor 312 may be coupled to the volatile memory 326 which may include Dynamic Random Access Memory (DRAM) and/or Static Random Access Memory (SRAM). The volatile memory 326 is typically large so that it can store dynamically loaded applications and data. As described further below, the volatile memory 326 may be configured in accordance with embodiments of the present invention.

The processor 312 may also be coupled to the memory device 12. The memory device 12 may include a read-only memory (ROM), such as an EPROM, and/or flash memory to be used in conjunction with the volatile memory 326. The size of the ROM is typically selected to be just large enough to store any necessary operating system, application programs, and fixed data. Additionally, the non-volatile memory 328 may include a high capacity memory such as a tape or disk drive memory.

The memory device 10 and volatile memory 326 may store various types of software, such as an operating system or office productivity suite including a word processing application, a spreadsheet application, an email application, and/or a database application.

While the invention may be susceptible to various modifications and alternative forms, specific embodiments have been shown by way of example in the drawings and have been described in detail herein. However, it should be understood that the invention is not intended to be limited to the particular forms disclosed. Rather, the invention is to cover all modifications, equivalents, and alternatives falling within the spirit and scope of the invention as defined by the following appended claims. 

1. A method comprising: generating a first value by sensing a data location at a first voltage; generating a second value by sensing the data location at a second voltage, wherein the second voltage is different from the first voltage; and combining the first value with the second value to read data conveyed by the data location.
 2. The method of claim 1, wherein combining comprises calculating a difference between the first value and the second value, calculating a ratio of the first value and the second value, calculating a sum of the first value and the second value, calculating a product of the first value and the second value, or a combination thereof.
 3. The method of claim 1, wherein generating a first value comprises counting a number of clock cycles that a data-stream has a value.
 4. The method of claim 1, wherein the second voltage is greater than the first voltage.
 5. The method of claim 1, wherein the first value is generated consecutive to the generation of the second value.
 6. The method of claim 1, wherein combining the first value with the second value comprises multiplying the second value with a predetermined weighting value and adding the product to the first value.
 7. The method of claim 1, wherein combining the first value with the second value comprises applying the first value and the second value to a neural network.
 8. The method of claim 1, wherein combining the first value with the second value comprises applying the first value and the second value to a Bayesian network.
 9. A method comprising: receiving an analog signal from a data location; converting the analog signal to a digital signal using an analog-to-digital converter; transmitting the digital signal to a counter; counting a number of clock cycles that the digital signal is at a logic high value and producing a first count indicative of a value stored in the data location.
 10. The method, as set forth in claim 10, wherein receiving an analog signal comprises receiving an analog signal from a data location along a bit line, and further comprising comparing a voltage of the bit line to a reference voltage to produce the digital signal.
 11. The method, as set forth in claim 10, wherein the reference voltage is generally constant.
 12. The method, as set forth in claim 10, wherein the reference voltage varies, and further comprising counting a number of clock cycles that the digital signal is at a logic high value and producing a second count indicative of the value stored in the data location.
 13. The method, as set forth in claim 12, comprising receiving the first count and the second count at an interfuser and identifying the value stored in the data location, based on a combination of the first count and the second count.
 14. A method comprising: converting an analog signal from a bit line of an array to a digital signal using an analog-to-digital converter; receiving the digital signal at a counter and producing two more counts by the counter; and receiving the two or more counts at an interfuser, and reading data conveyed on the bit line based on the two or more counts utilizing the interfuser.
 15. The method, as set forth in claim 14, wherein converting comprises comparing a voltage of the bit line to a reference voltage to produce the digital signal.
 16. The method, as set forth in claim 15, wherein comparing comprises comparing a voltage of the bit line to a variable reference voltage to produce the digital signal.
 17. The method of claim 14, wherein reading comprises combining the two or more counts.
 18. The method of claim 17, wherein combining comprises calculating a difference between the two or more counts, calculating a ratio of the two or more counts, calculating a sum of the two or more counts, calculating a product of the two or more counts, or a combination thereof.
 19. The method of claim 14, wherein converting the analog signal and receiving the digital signal comprise: sensing a data location on the bit line using both a first reference voltage and a second reference voltage; and outputting both a first value and a second value corresponding to the first reference voltage and the second reference voltage, respectively; and.
 20. The method of claim 19, wherein reading the data comprises combining the first value with the second value using the interfuser. 